Apparatus and method for local oscillator calibration in mixer circuits

ABSTRACT

An apparatus and method for local oscillator calibration compensates for filter passband variation in a mixer circuit, such as a receiver circuit. The receiver includes at least a mixer circuit and a filter coupled to the output of the mixer. During operation, the mixer mixes an RF input signal with a first local oscillator (LO) signal to frequency translate a selected channel in the RF input signal into the passband of the filter. During a calibration mode, the RF input signal is disabled, and the first LO signal is injected into the filter input by leaking the first LO signal through the mixer circuit. The frequency of the LO signal is then swept over a frequency bandwidth that is sufficiently wide so that the actual passband is detected by measuring the signal amplitude at the output of the bandpass filter, thereby determining any variation in the passband of the filter from the expected passband. Once the actual passband is determined, then the frequency of the first local oscillator signal is adjusted or tuned to compensate for any frequency shift of the actual passband compared to the expected passband. Therefore, the selected channel is up-converted into the center of the actual passband of the bandpass filter and will not fall outside the passband. This enables the passband of the bandpass filter to be narrowed, as compared with conventional receivers that do not utilize this calibration procedure. For example, the bandpass filter can be narrowed to one or two channels wide.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention generally relates to tuner calibration, and morespecifically to calibrating a local oscillator signal to compensate forvariation in the filter passband response of a channel selection filterin a dual conversion tuner.

2. Background Art

Television signals are transmitted at radio frequencies (RF) usingterrestrial, cable, or satellite transmission schemes. Terrestrial andcable TV signals are typically transmitted at frequencies ofapproximately 57 to 860 MHZ, with 6 MHZ channel spacings in the UnitedStates and 8 MHz channel spacing in Europe. Satellite TV signals aretypically transmitted at frequencies of approximately 980 to 2180 MHz.

Regardless of the transmission scheme, a tuner is utilized to select anddown-convert a desired channel from the TV signal to an intermediatefrequency (IF) signal or a baseband signal, which is suitable forprocessing and display on a TV or computer screen. The tuner shouldprovide sufficient image rejection and channel selection duringdown-conversion as is necessary for the specific application. TheNational Television Standards Committee (NTSC) sets standards fortelevision signal transmission, reception, and display. To process aNTSC signal, it is preferable that the tuner have a high-level of imagerejection. However, less image rejection is acceptable for non-NTSCsignals depending on the specific application and the correspondingdisplay requirements.

To achieve a high level of image rejection, traditional TV tunersutilize a dual-conversion architecture having two mixers and at leastone surface acoustic wave (SAW) filter. The first mixer up-converts thereceived RF signal to a first IF frequency (e.g. 1200 MHZ) that is fixedabove the RF signal band of the incoming TV signal, using a variablelocal oscillator (LO) signal. A SAW filter, centered at the first IF,selects the channel of interest and provides the image rejection toprevent signal interference. The second mixer then down-converts thefirst IF to a lower frequency second IF, using a second fixed frequencyLO signal. The second IF output is at baseband for a NTSC compatiblesignal. Alternatively, the second IF is at 36 or 44 MHZ for a cablesystem output that is fed into a set-top box or a cable modem. Channelselection is realized by adjusting the first LO signal so that thedesired channel is up-converted into the passband of the SAW filter, andis then down-converted to baseband by the second mixer.

The accuracy of the channel selection in the dual conversion tuner isdependent on the accuracy of the passband of the SAW filter. If thepassband of the SAW filter varies because of manufacturing tolerances,temperature variations, etc., then the accuracy of the channel selectionwill suffer. For example, if the passband varies from that intended,then a portion or all of the desired channel may fall outside the SAWpassband, causing unwanted signal attenuation in the desired channel.

A conventional method to address the passband tolerance of the SAWfilter is to simply increase the passband so as to pass a larger numberof channels than is necessary. For instance, SAW filters for TV tunerscan be designed to have a passband of 4 or more channels, so ascompensate for variation in the passband tolerance. The larger SAWpassband improves the likelihood that the desired channel will beup-converted into the SAW filter passband, but also means that one ormore undesired channels will also be passed. These unwanted channels cancause signal distortion in the down-conversion stage that requiresadditional filtering at baseband to correct.

What is needed is a method or apparatus for calibrating the dualconversion tuner (or other type of receiver) for the passband tolerancesof the SAW filter, so that the passband can be narrowed to passapproximately only 1 or 2 channels.

BRIEF SUMMARY OF THE INVENTION

The present invention is an apparatus and method for local oscillatorcalibration in mixer circuits. For example, a dual conversion receivercan include a first mixer, a second mixer, and a bandpass filter coupledbetween the first mixer and the second mixer. The dual conversionreceiver receives an RF input signal having a plurality of channels anddown-converts a selected channel to baseband or to a low frequency IFsignal.

During operation of a dual conversion receiver, the first mixer mixesthe RF input signal with a first local oscillator signal to up-convertthe RF input signal and generate a first IF signal. The bandpass filterselects a desired channel from the first IF signal that is within itsnarrow passband window, and substantially rejects all of the remainingchannels. Therefore, a particular channel is selected by varying thefrequency of the first local oscillator signal so that the desiredchannel is up-converted into the narrow passband of the bandpass filter.The second mixer then mixes the output of the bandpass filter with asecond local oscillator signal to down-convert the selected channel tobaseband, or to a low frequency second IF signal.

During a calibration mode, the RF input signal is disabled so that theactual passband of the bandpass filter can be determined using the firstlocal oscillator signal. The first local oscillator signal is injectedinto the input port of the bandpass filter either directly orindirectly. Since, the first local oscillator signal is coupled to thefirst mixer, the first local oscillator signal will leak through thefirst mixer to the input port of the bandpass filter when the RF signalis disabled. Therefore, the first local oscillator signal can be sweptover frequency and the actual passband of the bandpass filter can bedetermined by measuring the signal level output of the bandpass filter.The actual passband may differ from the expected passband due tomanufacturing tolerances or changes caused by temperature variation. Inother words, the passband may be shifted in frequency from that whichwas expected.

Once the actual passband is determined, then the frequency of the firstlocal oscillator signal is adjusted or tuned to compensate for anyfrequency shift of the actual passband verses the initial expectation.Therefore, the selected channel is up-converted into the center of theactual passband of the bandpass filter and will not fall outside thepassband. This enables the passband of the bandpass filter to benarrowed, as compared with conventional receivers that do not utilizethis calibration procedure. For example, the bandpass filter can benarrowed to one or two channels wide.

BRIEF DESCRIPTION OF THE DRAWINGS

The present invention is described with reference to the accompanyingdrawings. In the drawings, like reference numbers indicate identical orfunctionally similar elements. Additionally, the left-most digit(s) of areference number identifies the drawing in which the reference numberfirst appears.

FIG. 1A illustrates a dual conversion tuner with local oscillatorcalibration to compensate for filter passband variation.

FIG. 1B further illustrates the channel selection of the dual conversiontuner.

FIG. 1C illustrates a dual conversion tuner with local oscillatorcalibration and including image rejection.

FIG. 2 illustrates filter passband variation.

FIG. 3 illustrates a tuner calibration method for compensating forfilter passband variation.

FIG. 4 illustrates a filter characterization method using localoscillator signal injection.

FIG. 5 illustrates actual and expected filter passbands.

FIG. 6 illustrates a single stage mixer circuit having local oscillatorcalibration to compensate for filter passband variation.

DETAILED DESCRIPTION OF THE INVENTION

FIG. 1A illustrates a schematic of a tuner assembly 100 that has anautomatic gain control circuit (AGC) 102 and a tuner 134 that includes afilter calibration apparatus and method for detecting and compensatingfor filter passband variation.

The tuner assembly 100 receives an RF input signal 101 having multiplechannels and down-converts a selected channel to an IF frequency, toproduce an IF signal 133. For instance, the RF input signal 101 caninclude multiple TV channels that typically have 6 MHZ frequencyspacings and cover a range of 54-860 MHZ, and where the selected channelis down-converted to an IF frequency at 44 MHZ, 36 MHZ or some otherdesired IF frequency for further processing. The structure and operationof the tuner assembly 100 are described in further detail below.

The AGC circuit 102 provides automatic gain control using a variableresistor 104 and a low noise amplifier (LNA) 106. The variable resistor104 attenuates the RF input signal 101 according to a control signal103. In embodiments, the control signal 103 is based on the signalamplitude of the IF signal 133 so that the RF front-end gain can beadjusted to achieve a desired amplitude for the IF signal 133. The LNA106 provides low noise amplification and converts a single-ended inputsignal to a differential RF signal 107.

The tuner 134 has a dual conversion architecture (one up-conversion, andone down-conversion) that includes an up-convert mixer 108 and adown-convert mixer 118. The up-convert mixer 108 is driven by a firstphase locked loop (PLL) 110 that has coarse tuning capability from1270-2080 MHz. The down-convert mixer 118 is driven by a second PLL 124that has a relatively fixed frequency of 1176 MHZ (for a 44 MHZ IF) andhas fine frequency tuning capability. Two separate off-chip surfaceacoustic wave (SAW) filters 114 and 130 are used to perform IF filteringin the tuner 134. However, other bandpass filters besides SAW filterscould be used for the filters 114 and 130 as will be understood by thoseskilled in the arts. The first SAW filter 114 is connected between theup-convert mixer 108 and the down-convert mixer 118. The passband of theSAW filter 114 is centered at 1220 MHZ, and is preferably only a fewchannels wide (e.g. 1-2 channels wide or 12 MHZ for 6 MHZ TV channelspacings), and can be referred to as a channel selection filter. Thesecond SAW filter 130 has a passband at 44 MHZ and is coupled to theoutput of the amplifier 128. Additionally, various on-chip amplifiers112, 112, 128, and 132 are included throughout the tuner 134 to providesignal amplification, as necessary. The amplifier 116 is a variable gainamplifier controlled by a power detector 136 so as to provide automaticgain control as will be discussed further below. The power detector 136can also be called a signal detector or just a detector, and the VGA 116and the power detector 136 can be referred to as an automatic gaincontrol.

The operation of the tuner 134 is described as follows and in referenceto FIG. 1B, where FIG. 1B represents the frequency spectrum of theparticular signals that are operated on and generated by the tuner 134.The up-convert mixer 108 mixes the RF signal 107 with a LO signal 109that is generated by the PLL 110. As discussed above and as shown inFIG. 1B, the RF signal 107 can be a TV signal having a plurality ofchannels that occupy from 54 MHz to 860 MHz. Since the PLL 110 istunable from 1270-2080 MHZ, the RF signal 107 is up-converted to a firstIF 111 having a frequency that is above the 54-860 MHZ input frequencyband. The first IF 111 is sent off-chip to the SAW filter 114, which hasa narrow passband window centered at 1220 MHz, as discussed above. Thefirst SAW filter 114 selects a desired channel 115 that is within itsnarrow passband window, and substantially rejects all of the remainingchannels. Therefore, a particular channel is selected by varying thefrequency of the LO signal 109 so that the desired channel isup-converted into the narrow passband of the IF filter 114. The desiredchannel 115 (at 1220 MHZ) is sent back on-chip to the PGA 116, where thePGA 116 and the power detector 136 provide automatic gain control forthe selected channel 115. The down-convert mixer 118 mixes the output ofthe PGA 116 with an LO signal from the PLL 124. The down-convert mixer118 down-converts the desired channel 115 to an 44 MHZ IF signal 127that appears at the IF output of the down-convert mixer 118. Finally,the IF signal 127 is filtered a second time by the bandpass SAW filter130 to reject any unwanted frequency harmonics, producing the output IFsignal 133 at 44 MHZ, or some other desired IF frequency or baseband,and carrying the information in the desired channel.

In one embodiment, the down-conversion mixer 118 is an image rejectionmixer as shown in FIG. 1C. The image rejection mixer 118 includes twocomponent mixers 120 a and 120 b and a polyphase filter 126, where thecomponent mixers 120 a and 120 b are driven by a quadrature LO signal119 from apolyphase filter 122. The image rejection mixer 118down-converts the desired channel 116 to the IF signal 127 that appearsat the output of the polyphase filter 126, where the I and Q componentsof the IF signal 127 are combined in the polyphase filter 126 to provideimage rejection.

The specific frequencies mentioned in the description of the tunerassembly 100, and throughout this application, are given for examplepurposes only and are not meant to be limiting. Those skilled in thearts will recognize other frequency applications for the tuner assembly100 based on the discussion given herein. These other frequencyapplications are within the scope and spirit of the present invention.

Furthermore, the present invention is not limited to the dual conversiontuner 100, but can be utilized with any mixer or receiver circuit thatcan benefit from filter characterization and compensation. For instance,the present invention could be implemented with a direct conversiontuner having a single mixer circuit and a filter afterwards.

Furthermore, it is noted that the tuner 100 is configured fordifferential operation. For instance, the first mixer 108, the bandpassfilter 114, the second mixer 118, the first LO signal 109, and thesecond LO signal 119 are all configured with differential inputs andoutputs to reduce signal distortion. However, the present invention isnot limited to differential operation, and can be implemented in singleended configurations.

As discussed above, the filters 114 and 130 are subject to manufacturingtolerances and temperature variations that can cause their respectivepassbands to shift over frequency. For example, FIG. 2 illustrates adesired (or expected) passband 204 for the filter 114 that is centeredon 1220 MHz. However, the actual passband may vary from part-to-part andover temperature, so that the passband is shifted in frequency asrepresented by the actual passbands 202 and 206. Since the filterpassband is only a few channels wide, a small frequency shift can causethe channel at 1220 Mhz to be unexpectedly attenuated if a portion (orall) of the desired channel falls outside the passband of the filter.

As discussed above, the conventional solution used to address the filtertolerance is to simply widen the filter passband, improving the chancesthat the desired channel at 1220 MHz will pass unattenuated. However,widening the filter passband also passes more undesired channels thatcan cause distortion and interference in the second down-conversionstage. Accordingly, the present invention characterizes the actualpassband of the filter 114 in a calibration period (or mode) prior tochannel tuning and down-conversion. In other words, the passband of thefilter 114 is characterized with no input RF signal 101 during thecalibration mode, so as to detect any passband variations compared tothat which was expected (e.g. 1220 MHz in this example). Once the filterpassband is characterized, the first PLL 110 is tuned so the desiredchannel in the IF signal 111 is up-converted to the center of the actualpassband of the filter 114, which may or may not be at 1220 Mhz in thisexample. In other words, if the passband of the filter 114 is frequencyshifted from that which was expected, then the PLL 110 is tuned so thatthe first IF signal 111 compensates for the frequency shift of thefilter 114 passband. The filter characterization and compensation in adual conversion tuner is discussed further below using the flowchart 300that is shown in FIG. 3.

In step 302, the RF input signal 101 is disabled for the calibrationmode. For example, the filter characterization is performed when the RFinput signal 101 is not present or is disconnected for channel tuningand selection.

In step 304, the actual passband of the filter 114 is characterizedusing a calibration signal, such as the first local signal 109. Theactual passband of the filter 114 may be stored in a memory device (notshown). The details of the filter characterization will be discussedfurther with respect to the flowchart 400.

In step 306, the RF input signal 101 is enabled and a selected channelis identified for channel selection and down-conversion. For example,the a LO control 138 can receive a control signal 140 that identifies achannel selection for down-conversion.

In step 308, the frequency of the first local oscillator signal 109 isdetermined to frequency shift the desired channel into the actualpassband of the bandpass filter 114, and includes adjustments toaccommodate for any variations in the actual filter passband from theexpected filter passband. In other words, the first local oscillatorsignal is adjusted so that the selected channel is up-converted to thecenter frequency of the actual passband of the filter 114 by the mixer108. For instance, assuming the desired channel is at 50 Mhz in the RFsignal 101, the PLL 110 would be tuned to generate a local oscillatorsignal at 1270 MHz, to up-convert the selected channel to a first IFsignal of 1220 MHz. However, if the actual passband of the filter 114was found to be centered at 1222 MHz, then the PLL 110 would be tuned togenerate a local oscillator signal 109 at 1272 MHz. In other words, thefrequency of the first local oscillator 109 is increased or decreased sothat the desired channel is up-converted to the center of the actualpassband of the filter 114. Summarized another way, the frequency of thefirst local oscillator signal is adjusted to accommodate for the actualpassband of the filter 114, including tolerance variation that cause thefilter passband to change verses the expected passband.

In step 310, the selected channel in the RF input signal is up-convertedto the center of the actual passband of the filter 114 using the localoscillator signal 109 that was generated in step 308. For example, themixer 108 up-converts the selected channel in the RF input signal 101 tothe center of the actual passband of the filter 114.

The filter characterization of step 304 is discussed further below usingthe flowchart 400 that is shown in FIG.4. The filter 114 passband ischaracterized by leaking the LO signal 109 through the filter 114 whenthere is no RF signal input 101. The frequency of the LO signal 109 isvaried or swept over frequency so as to trace-out the frequency passbandof the filter 114.

In step 402, LO signal 109 is injected into the input of the filter 114so as to characterize the actual passband of the filter. For example,the mixer 108 will typically leak a portion of the local oscillatorsignal 109 to its IF port that is connected to the input of the filter114, where the leakage is characterized by the LO-to-IF isolation.Therefore, the mixer LO leakage can be utilized to inject the firstlocal oscillator signal 109 into the input of the mixer 108.Alternatively, the LO signal 109 can be directly injected into the inputof filter 114 using a switch and signal path (not shown) that bypassesthe mixer 109 and connects to the input port of the filter 114.

In step 404, the output of the filter 114 is detected or measuredresponsive to the LO signal 109 that is injected into the input of thefilter 114. For example, the power detector 136 can be used to monitorthe signal amplitude or power output of the filter 114 responsive to theLO signal 109, the result of which is forwarded to the LO control 138.As discussed above, the power detector 136 also provides feedback forautomatic gain control of the programmable gain amplifier 116 signal. Inother words, portions of the automatic gain control (AGC) can also beused to measure the signal level at the output of the bandpass filter114.

In step 406, the frequency of the LO signal 109 is varied to sweepacross the actual passband of the filter 114, and steps 402 and 404 arerepeated above to trace out the actual passband of the filter asrepresented by FIG. 5. For instance, the frequency of the LO signal 109can be swept across the expected passband of the filter, and beyond todetect any shifts in the passband, to trace out the actual passband ofthe filter.

In step 408, the LO control 138 receives the amplitude output (or signallevel output) from the power detector 136 that reflects the actualpassband of the filter 114. The LO control 138 determines anyadjustments to the PLL 110 as are necessary so that the selected channelis up-converted to the center of the actual passband of the filter 114.In other words, as described in step 308, the frequency of the localoscillator signal 109 is adjusted to account for any shift in frequencyof the actual filter 114 passband from the ideal so that the selectedchannel can be up-converted to the actual filter passband 202 that isshown in FIG. 4. For instance if the selected channel is at 100 MHz inthe RF input signal 101, then the LO signal 109 should be set to 1320MHz, if the filter 114 has a passband that is centered on the idealvalue of 1220 MHz. However, if the actual filter passband of the filter114 is shifted to say 1215 MHz as shown in FIG. 4, then the LO signal109 should be set to 1315 MHz to up-convert the 100 MHz channel to 1215MHz. In this manner, the frequency shift of the actual filter 114passband from expected is accommodated and compensated.

As discussed above, the actual filter bandwidth of the filter 114 isdetermined by leaking the LO signal 109 in the input of the filter 114,and detecting the filter output with the power detector 136, so as tocharacterize the actual filter passband of the filter. This may bereferred to as a calibration mode since the RF input signal is disabledduring this period of filter characterization. Afterwhich, duringchannel selection and down-conversion, the frequency of the LO signal isadjusted to account for any variation in the actual filter passband fromthe expected filter passband.

Since the actual filter passband of the filter 114 is determined, thepassband of the filter can be significantly narrowed to pass only 1 or 2channels. This is advantageous because it reduces the number of unwantedchannels that are down-converted, thereby reducing distortion andinterference during the second down-conversion step.

The invention herein has been described in reference to a dualconversion tuner for the down-conversion and processing of televisionsignals. However, the invention is not limited to this exampleembodiment, and could be implemented in any receiver. More specifically,the invention could be implemented in any receiver having a filter andlocal oscillator that could benefit from filter characterization using alocal oscillator signal or other calibration signal, and thencompensating for any filter variation using the local oscillator signal.For example, FIG. 6 illustrates a single stage mixer circuit 600 havingonly the first stage components from the dual conversion receiver 100 ofFIG. 1A. As discussed above, the LO 110 is swept over frequency during acalibration mode, and the output of the filter 114 is measured todetermine the actual passband of the filter 114. Afterwhich, during IFprocessing, the LO 110 is tuned so as to compensate for any shift in theactual passband of the filter 114 from the expected passband. In otherwords, the LO 110 is tuned so the desired portion of the RF input signalis frequency converted to an IF frequency that falls in the actualpassband of the filter 114, despite any passband variation due totemperature drift, manufacturing tolerances, etc.

Conclusion

Example embodiments of the methods, systems, and components of thepresent invention have been described herein. As noted elsewhere, theseexample embodiments have been described for illustrative purposes only,and are not limiting. Other embodiments are possible and are covered bythe invention. Such other embodiments will be apparent to personsskilled in the relevant art(s) based on the teachings contained herein.Thus, the breadth and scope of the present invention should not belimited by any of the above-described exemplary embodiments, but shouldbe defined only in accordance with the following claims and theirequivalents.

1. In a receiver having a first mixer, a second mixer, and a bandpassfilter coupled between said first mixer and said second mixer, saidfirst mixer responsive to a first local oscillator signal that iscoupled to said first mixer and said second mixer responsive to a secondlocal oscillator signal, a method of compensating for passband variationof said bandpass filter, comprising: injecting said first localoscillator signal into an input port of said bandpass filter;determining an actual passband of said bandpass filter responsive tosaid first local oscillator signal; mixing an RF input signal havingplurality of channels with said first local oscillator signal after saidstep of determining to generate a first IF signal, including said stepof adjusting a frequency of said first local oscillator signal basedupon a selected channel of said plurality of channels and based uponsaid actual passband of said bandpass filter.
 2. The method of claim 1,wherein the step of determining includes the steps of: sweeping saidfrequency said first local oscillator signal; and measuring an output ofsaid bandpass filter responsive said sweeping step, to determine saidactual passband of said bandpass filter.
 3. The method of claim 1,wherein the step of adjusting said frequency of said first localoscillator signal includes the step of setting a frequency of said firstlocal oscillator signal so said selected channel in said first IF signalfalls within said actual passband of said bandpass filter.
 4. The methodof claim 1, wherein said step of adjusting said frequency includes thestep of setting said frequency of said first local oscillator signal soas to compensate for variation of said actual passband of said bandpassfilter.
 5. The method of claim 4, wherein said variation is caused by atemperature variation of said bandpass filter.
 6. The method of claim 4,wherein said variation is caused by manufacturing tolerance variation ofsaid bandpass filter.
 7. The method of claim 1, wherein said of step ofinjecting includes the steps of: coupling said first local oscillatorsignal to a local oscillator port of said first mixer when said RF inputsignal is disabled.
 8. The method of claim 7, further comprising thestep of leaking said first local oscillator signal through said firstmixer to an input port of said bandpass filter.
 9. The method of claim1, wherein said step of mixing includes the step of up-converting saidselected channel in said first IF signal into said actual passband ofsaid bandpass filter.
 10. The method of claim 9, further comprising thestep of filtering said first IF signal so that said selected channel andat most one other channel pass through said bandpass filter.
 11. Themethod of claim 9, further comprising the step of filtering said firstIF signal so that only said selected channel passes through saidbandpass filter.
 12. The method of claim 9, further comprising the stepof mixing said selected channel at an output of said bandpass filterwith a second local oscillator signal in said second mixer todown-converted said selected channel to baseband.
 13. The method ofclaim 9, wherein said step of mixing said selected channel includes thestep of providing image rejection for said selected channel.
 14. Themethod of claim 1, further comprising the step of filtering said firstIF signal to generate an output passband that passes said selectedchannel and at most one other channel.
 15. A receiver for processing anRF input signal having a plurality of channels, comprising: a receiverinput configured to receive an RF input signal having a plurality ofchannels; a first mixer having a first input coupled to said receiverinput and a second input coupled to a first local oscillator signal; abandpass filter having a passband and an input coupled to an IF outputof said first mixer; and a second mixer having an first input coupled anoutput of the bandpass filter and an second input coupled to a secondlocal oscillator signal; wherein said passband of said bandpass filteris determined by sweeping a frequency of said first local oscillatorsignal during a calibration mode.
 16. The receiver of claim 15, whereinafter said calibration mode, said frequency of said first localoscillator is adjusted so that a selected channel of said plurality ofchannels falls in said passband of said bandpass filter that isdetermined during said calibration mode.
 17. The receiver of claim 15,wherein after said calibration mode, said frequency of said first localoscillator signal is adjusted to account for any passband variation sothat said selected channel of said plurality of selected channels isup-converted into said passband of bandpass filter
 18. The receiver ofclaim 15, further comprising a means for detecting a power output ofsaid bandpass filter responsive to said first local oscillator duringsaid calibration mode, said passband determined from said power output.19. The receiver of claim 18, further comprising a local oscillatorcontrol module that receives said power output from said bandpass filterand determines said passband of said bandpass filter based on said poweroutput, and controls a frequency of said first local oscillator signalresponsive to said passband of said bandpass filter.
 20. The receiver ofclaim 15, wherein said RF input signal is disabled during saidcalibration mode.
 21. The receiver of claim 15, wherein during saidcalibration mode, said local oscillator signal is swept over a frequencybandwidth sufficient to include said passband of said bandpass filter.22. The receiver of claim 15, wherein during said calibration mode, saidlocal oscillator signal is swept from a first frequency to a secondfrequency, said passband of said bandpass filter within a bandwidthdefined by said first frequency and said second frequency.
 23. Thereceiver of claim 15, wherein at least one of said first mixer, saidsecond mixer, and said bandpass filter includes differential inputs anddifferential outputs.
 24. The receiver of claim 15, wherein said firstmixer, said second mixer, and said bandpass filter are differential. 25.The receiver of claim 15, wherein said first mixer and said second mixerare disposed on a common substrate.
 26. The receiver of claim 25,wherein said bandpass filter is disposed external to said commonsubstrate.
 27. A receiver for processing an RF input signal having aplurality of channels, comprising: a receiver input configured toreceive said RF input signal having said plurality of channels; a firstmixer having a first input coupled to said receiver input and a secondinput coupled to a first local oscillator signal; a bandpass filterhaving passband and an input coupled to an IF output of said firstmixer; a programmable gain amplifier (PGA) having an input coupled to anoutput of said bandpass filter; a second mixer having a first inputcoupled to an output of said programmable gain amplifier and an secondinput coupled to a second local oscillator signal; a detector circuitthat detects a signal level at an output of said PGA and controls a gainof said PGA based on said signal level; an LO control circuit thatadjusts a frequency of said first local oscillator signal based on (1) aselected channel of said plurality of channels, and (2) a passband ofsaid bandpass filter determined during a calibration mode.
 28. Thereceiver of claim 27, wherein said frequency of said first localoscillator signal is swept during said calibration mode, and saiddetector circuit detects said signal level at said output of said PGAresponsive to said first local oscillator signal to determine saidpassband of said bandpass filter.
 29. The receiver of claim 27, whereinsaid LO control circuit adjusts said frequency of said first localoscillator signal so that said selected channel of said plurality ofchannels falls in said passband of said bandpass filter.
 30. Thereceiver of claim 27, wherein said passband of said bandpass filter isat most 2 channels wide.
 31. The receiver of claim 27, wherein saidfirst mixer and said second mixer are differential.
 32. The receiver ofclaim 31, wherein said first mixer and said second mixer are disposed ona common integrated circuit substrate, and said bandpass filter isdisposed external to said common integrated circuit substrate.
 33. Thereceiver of claim 31, wherein said second mixer is an image rejectionmixer.
 34. A receiver for processing an RF input signal having aplurality of channels, comprising: a receiver input configured toreceive said RF input signal having said plurality of channels; a firstmixer having a first input coupled to said receiver input and a secondinput coupled to a first local oscillator signal, wherein a frequency ofsaid first local oscillator signal determines a channel selection forsaid receiver; a bandpass filter having a passband and an input coupledto an IF output of said first mixer; a second mixer having an firstinput coupled an output of said bandpass filter and an second inputcoupled to a second local oscillator signal; means for determining atolerance variation of said passband of said first bandpass filter; andmeans for adjusting said frequency of said first local oscillator signalso as to compensate for said tolerance variation of said passband ofsaid bandpass filter.
 35. The receiver of claim 34, wherein saidtolerance variation of said passband of said bandpass filter is a due totemperature variation.
 36. The receiver of claim 34, wherein saidtolerance variation of said passband of said bandpass filter is due to amanufacturing variation.